Dual-path amplifier having reduced harmonic distortion

ABSTRACT

An embodiment of a dual-path amplifier includes a power splitter connected to first and second power amplifiers respectively connected to first and second transmission lines connected to a power combiner having a phase-offset deficit at the second harmonic frequency 2f0, where the first and second transmission lines are designed to provide a complementary phase offset at 2f0 substantially equal to the phase-offset deficit such that the two amplified signals will be combined at the power converter with a total phase offset at 2f0 of about 180 degrees in order to reduce harmonic distortion in the amplified output signal, without substantially diminishing the output power at the fundamental frequency f0. In certain PCB-based implementations, the transmission lines include metal traces and lumped elements providing different impedance transformations that achieve the complementary phase offset, where the metal traces may have significantly different physical and electrical characteristics.

TECHNICAL FIELD

Embodiments of the subject matter described herein relate generally toamplifiers, and more particularly to dual-path power amplifiers havingtwo parallel amplification paths.

BACKGROUND

Wireless communication systems employ power amplifiers for increasingthe power of radio frequency (RF) signals. In a cellular base station,for example, a dual-path power amplifier may form a portion of the lastamplification stage in a transmission chain before provision of theamplified signal to an antenna for radiation over the air interface.High gain, high linearity, stability, and a high level of power-addedefficiency are characteristics of a desirable power amplifier in such awireless communication system.

FIG. 1 is a simplified schematic circuit diagram of a conventionaldual-path Class AB power amplifier 100 having a power splitter 120, twoparallel amplification paths 130(1) and 130(2), and a power combiner140, where each amplification path 130(1), 130(2) has an amplifier132(1), 132(2) and a transmission line 136(1), 136(2) that connects theoutput of the amplifier 132(1), 132(2) to the power combiner 140. Thoseskilled in the art will understand that real-world implementations ofthe dual-path amplifier 100 may have additional components that are notshown in FIG. 1.

The power splitter 120 divides an input signal 112 received at inputnode 110 into two different signal components 122(1) and 122(2). In someimplementations, the power splitter 120 is a 50-50 splitter that dividesthe input signal 112 into two components 122(1) and 122(2) havingsubstantially equal power levels.

Each amplifier 132(1), 132(2) amplifies its corresponding component122(1), 122(2) to generate a corresponding amplified component 134(1),134(2) that gets injected into the corresponding transmission line136(1), 136(2). As understood by those skilled in the art, theelectrical characteristics of each transmission line 136(1), 136(2)result in some degree of attenuation and some amount of phase shift asthe corresponding amplified component 134(1), 134(2) traverses thetransmission line 136(1), 136(2) towards the power combiner 140.

The power combiner 140 receives and combines the components 138(1) and138(2) output from the two respective transmission lines 136(1) and136(2) to generate an amplified output signal 142 at the output node150. In order to optimize the power of the amplified output signal 142,in conventional, real-world implementations, the two transmission lines136(1) and 136(2) are specifically designed to have substantiallyidentical electrical characteristics at the fundamental frequency ofoperation, f0, of the amplifier 100, so that the two components 138(1)and 138(2) will be combined at the power combiner 140 with minimallosses at that fundamental frequency.

FIG. 2 is a simplified, top view of a conventional, real-worldimplementation 200 of the dual-path amplifier 100 of FIG. 1 on a printedcircuit board (PCB) 205. Mounted onto the PCB 205 is a packagedintegrated circuit (IC) device 232 that contains the two amplifiers132(1) and 132(2) of FIG. 1. The two transmission lines 136(1) and136(2) of FIG. 1 are implemented using two respective metal traces236(1) and 236(2) on the PCB 205.

According to a known technique, the power splitter 120 and the powercombiner 140 of FIG. 1 are implemented as two instances 220 and 240 ofthe same discrete, hybrid coupler that are mounted onto the PCB 205 andrespectively configured to operate symmetrically as a power splitter anda power combiner. In a typical conventional implementation, the hybridcoupler has been specifically designed to operate optimally at therelevant fundamental frequency, f0. As known in the art, when configuredas the power splitter 220, the first instance of the hybrid coupleroutputs the two signal components 122(1) and 122(2) having a phaseoffset substantially equal to 90 degrees at the fundamental frequency.Similarly, when configured as the power combiner 240, the secondinstance of that same hybrid coupler applies a symmetric phase offsetsubstantially equal to 90 degrees between the two amplified signalcomponents 138(1) and 138(2) at the fundamental frequency as thecomponents are being combined such that the combined effect of the twosymmetric 90-degree phase offsets results in (substantially) zero netphase offset at the fundamental frequency when the two components arecombined. This zero net phase offset at the fundamental frequency isachieved by applying the 90-degree phase offset symmetrically to eachpath: one at the power splitter 220 and the other at the power combiner240.

In such a conventional implementation, the physical layouts of the twometal traces 236(1) and 236(2) are specifically designed to be mirrorimages of each other, such that the two metal traces will havesubstantially identical electrical characteristics at the fundamentalfrequency. Accordingly, the respective phase shifts applied by the twometal traces 236(1) and 236(2) to the two amplified components 134(1)and 134(2) output from the amplifiers 132(1) and 132(2) at thefundamental frequency will be substantially identical, so that theoptimal signal combining at the power combiner 240 at the fundamentalfrequency is not adversely impacted by the metal traces.

BRIEF DESCRIPTION OF THE DRAWINGS

A more complete understanding of the subject matter may be derived byreferring to the detailed description and claims when considered inconjunction with the following figures, wherein like reference numbersrefer to similar elements throughout the figures.

FIG. 1 is a simplified schematic circuit diagram of a conventionaldual-path Class AB power amplifier;

FIG. 2 is a simplified, top view of a conventional, real-worldimplementation of the dual-path amplifier of FIG. 1 on a printed circuitboard;

FIG. 3 is a simplified schematic circuit diagram of a dual-path Class ABpower amplifier, according to certain embodiments of the presentdisclosure;

FIG. 4 is a graphical representation of the trajectories of theimpedance transformations imposed by two example transmission lines ofFIG. 1 having the same voltage standing wave ratio (VSWR) at thefundamental frequency f0;

FIG. 5 is a graphical representation of the trajectories of theimpedance transformations imposed by another two example transmissionlines of FIG. 1 having the same VSWR at f0;

FIG. 6 is a graphical representation of the trajectories of theimpedance transformations imposed by two example transmission lines ofFIG. 3;

FIG. 7 is a flow diagram of the process for implementing thetransmission lines of FIG. 3 on a printed circuit board according to oneembodiment of the disclosure; and

FIG. 8 is a simplified, top view of a PCB-based implementation of thedual-path amplifier of FIG. 3, according to certain embodiments of thepresent disclosure.

DETAILED DESCRIPTION

As described above, in conventional practice, the metal traces 236(1)and 236(2) in the PCB-based implementation 200 of FIG. 2 for thedual-path amplifier 100 of FIG. 1 are intentionally designed to havephysical shapes that are mirror images of one another such that themetal traces have substantially identical electrical characteristics atthe fundamental frequency of operation (f0) of the amplifier 100. Such adesign achieves optimal combining of the two amplified signal components138(1) and 138(2) at the fundamental frequency. However, such a designstrategy ignores the distortion that can occur in the output signal 142at frequencies other than the fundamental frequency, for example, at thesecond harmonic frequency (i.e., twice the fundamental frequency or2f0).

In an ideal implementation, a hybrid coupler having a 90-degree phaseoffset at the fundamental frequency (f0) would have a 180-degree phaseoffset at the second harmonic frequency (2f0) corresponding to thatfundamental frequency. In such an ideal implementation, the powersplitter 120 would output the two signal components 122(1) and 122(2)with a 90-degree phase offset at f0 and a 180-degree phase offset at2f0. Furthermore, the power combiner 140 would ideally apply acomplementary 90-degree phase offset at f0 and a complementary180-degree phase offset at 2f0.

Note that the amplifiers 132(1) and 132(2) operate to amplify the signalcomponents 122(1) and 122(2) at the fundamental frequency f0 (i.e.,signal energy at the fundamental frequency). Nevertheless, nonlinearamplification characteristics of the amplifiers 132(1) and 132(2) willintroduce signal distortion into the amplified signals 134(1) and 134(2)such that the amplified signals 134(1) and 134(2) output from theamplifiers 132(1) and 132(2) will contain amplified signal components atthe fundamental frequency f0 as well as harmonic distortion componentsof the fundamental frequency f0 at frequencies 2f0, 3f0, 4f0, etc. Inaddition, the electrical characteristics of the transmission lines134(1) and 134(2) can impact and increase the magnitude of the signalenergy at the second harmonic frequency 2f0 in the amplified signals138(1) and 138(2) that are applied to the hybrid power combiner 140 whenthe amplified signals 138(1) and 138(2) combine out of phase (i.e., witha phase offset of 180 degrees).

In an ideal implementation in which the power combiner 140 applies a90-degree phase offset at f0 and a 180-degree phase offset at 2f0, the180-degree phase offset at 2f0 will result in a destructive interferencebetween the two signals at 2f0, thereby reducing harmonic distortion inthe output signal 142.

In real-world implementations, however, a hybrid coupler having a90-degree phase offset at the fundamental frequency f0 will not have a180-degree phase offset at the second harmonic frequency 2f0. Forexample, a real-world hybrid coupler designed to have a 90-degree phaseoffset at f0, might have a phase offset at 2f0 of only about 135degrees. In that case, when the two amplified signals 138(1) and 138(2)are combined at the power combiner 140, the 135-degree phase offset willresult in imperfect cancellation of the signal energy at 2f0 leading toundesirable harmonic distortion in the output signal 142.

FIG. 3 is a simplified schematic circuit diagram of a dual-path Class ABpower amplifier 300, according to certain embodiments of the presentdisclosure. As shown in FIG. 3, the power amplifier 300 has a powersplitter 320, two parallel amplification paths 330(1) and 330(2), and apower combiner 340, where each amplification path 330(1), 330(2) has anamplifier 332(1), 332(2) and a transmission line 336(1), 336(2) thatconnects the output of the amplifier 332(1), 332(2) to the powercombiner 340. Those skilled in the art will understand that real-worldimplementations of the dual-path amplifier 300 may have additionalcomponents that are not shown in FIG. 3.

The power splitter 320 divides an input signal 312 received at its inputnode 310 into two different signal components 222(1) and 222(2) at itstwo output nodes. In some implementations, the power splitter 320 is a50-50 splitter that divides the input signal 312 into two components322(1) and 322(2) having substantially equal power levels.

Each amplifier 332(1), 332(2) amplifies its corresponding component322(1), 322(2) to generate a corresponding amplified component 334(1),334(2) that gets injected into the corresponding transmission line336(1), 336(2). As understood by those skilled in the art, theelectrical characteristics of each transmission line 336(1), 336(2)result in some degree of attenuation and some amount of phase shift asthe corresponding amplified component 334(1), 334(2) traverses thetransmission line 336(1), 336(2) towards the power combiner 340.

The power combiner 340 receives, at its two input ports, the components338(1) and 338(2) output from the two respective transmission lines336(1) and 336(2) to generate an amplified output signal 342 at itsoutput node 350.

As will be described in detail below, a significant difference betweenthe power amplifier 300 of FIG. 3 and the prior-art power amplifier 100of FIG. 1 is that the transmission lines 336(1) and 336(2) of FIG. 3 arespecifically designed to have different physical and electricalcharacteristics such that the phase offset between the two transmissionlines at the second harmonic frequency 2f0 substantially compensates fora non-zero phase-offset deficit of the power combiner 340 at 2f0 withoutsignificantly impacting the phase offset between the two transmissionlines at the fundamental frequency f0. In the prior-art power amplifier100 of FIG. 1, the transmission lines 136(1) and 136(2) are specificallydesigned to be mirror images of one another having identical electricalcharacteristics such that there is no substantial phase offset betweenthe two transmission lines at the fundamental frequency f0 without anyregard for the phase offset between the two transmission lines at thesecond harmonic frequency 2f0.

Those skilled in the art will understand that, when the power amplifier300 is implemented on a printed circuit board, each transmission line336(1), 336(2) may be implemented using various impedance elements thatprovide different combinations of inductance, capacitance, and/orresistance that contribute to the overall phase shift between the outputof the amplifier 332(1), 332(2) and the corresponding input to the powercombiner 340. For example, a segment of a metal trace formed on the PCBsubstrate may be used to provide inductance. Similarly, bond wires andother types of leads may also be used to provide inductance. Lumpedcapacitors connected in series or shunt configurations may be used toprovide capacitance. In one possible implementation, each transmissionline 336(1), 336(2) of FIG. 3 may be implemented as a PCB metal tracehaving a number of interconnected trace segments with a lumped capacitorconnected in a shunt configuration to one end of the metal trace, wherethe two transmission lines 336(1) and 336(2) have different electricallengths and different impedances at 2f0.

A goal of the present disclosure is to design the two transmission lines336(1) and 336(2) to have different electrical characteristics suchthat, at the second harmonic frequency 2f0, the phase offset resultingfrom traversing the transmission lines combined with the phase offsetinduced by the hybrid power combiner 340 will be close to 180 degreeswithout significantly affecting the phase match at the fundamentalfrequency f0, such that the signal components will substantiallydestructively interfere at 2f0 with minimal mismatch losses at f0. Whenthe transmission lines 336(1) and 336(2) are designed and the powercombiner 340 is configured such that their respective phase offsets arecumulative, those phase offsets are said to be complementary.

According to certain real-world implementations of the presentdisclosure, the transmission lines 336(1) and 336(2) of the dual-pathamplifier 300 of FIG. 3 are specifically designed to compensate for thedeficiency in the phase offset imposed by the power combiner 340 at thesecond harmonic frequency. The two transmission lines 336(1) and 336(2)can be practically implemented on a printed circuit board to have aphase offset up to about 80 degrees at 2f0. As long as the phase-offsetdeficit of the power combiner 340 at 2f0 is no more than about 80degrees, the transmission lines 336(1) and 336(2) can be implemented tohave a complementary phase offset that compensates for that phase-offsetdeficit.

Thus, for example, for a power combiner 340 that imposes a 135-degreephase offset at 2f0, the transmission lines 336(1) and 336(2) arespecifically designed to achieve a complementary phase offset of about45 degrees at 2f0 such that the total net phase offset at 2f0 will beabout 180 degrees so that significant harmonic cancellation will occurwhen the two amplified signals 338(1) and 338(2) are combined at thepower combiner 340. In addition, in order not to significantly adverselyaffect the constructive interference at the fundamental frequency f0,the transmission lines 336(1) and 336(2) are also specifically designedto impose a phase offset substantially close to zero at f0.

In general, for each transmission line 336(1), 336(2) the impedanceZload presented to the power combiner 340 of FIG. 3 is given by Equation(1) as follows:

$\begin{matrix}{{{Zload} = {{Zo}*\frac{{Zout} + {j*{Zo}*{{Tan}\left\lbrack {B} \right\rbrack}}}{{Zo} + {j*{Zout}*{{Tan}\left\lbrack {B} \right\rbrack}}}}},} & (1)\end{matrix}$

where:

Zo is the impedance of the corresponding transmission line 336(1),336(2);

Zout is impedance from the corresponding amplifier 332(1), 332(2) thatinjects an amplified signal 334(1), 334(2) into the transmission line336(1), 336(2); and

Bl is the phase length of the transmission line 336(1), 336(2).

In order to achieve substantially uniform performance across thefrequency band of interest about the fundamental frequency f0, eachtransmission line 336(1), 336(2) should have the same amount of mismatchloss (in dB) at f0, which can be achieved by designing both transmissionlines 336(1) and 336(2) to have the same complex power reflectioncoefficient Γ. This can be accomplished by designing the transmissionlines 336(1) and 336(2) to have the same voltage standing wave ratio(VSWR) given by Equation (2) as follows:

$\begin{matrix}{{VSWR} = \frac{\left( {1 + \Gamma} \right)}{\left( {1 - \Gamma} \right)}} & (2) \\{\Gamma = {{\Gamma_{Re} + {i\;\Gamma_{Im}}} = \frac{\left( {{Zload} - {ZoComb}} \right)}{\left( {{Zload} + {ZoComb}} \right)}}} & (3) \\{\rho = \sqrt{\Gamma_{Re}^{2} + \Gamma_{Im}^{2}}} & (4) \\{\theta = {\arctan\left( \frac{\Gamma_{Im}}{\Gamma_{Re}} \right)}} & (5)\end{matrix}$

Γ_(Re) and Γ_(Im) are the real and imaginary components of the complexpower reflection coefficient Γ, ZoComb is the impedance from the powercombiner 340, ρ is the magnitude of the power reflection coefficient Γ,and θ is the phase angle of the power reflection coefficient Γ.

Equation (3) can be re-written as Equation (6) as follows:

$\begin{matrix}{{Zload} = {{ZoComb}\frac{\left( {1 + \Gamma} \right)}{\left( {1 - \Gamma} \right)}}} & (6)\end{matrix}$

The phase shift Ø of a transmission line 336(1), 336(2) is given byEquation (7) as follows:

$\begin{matrix}{\varnothing = {\arctan\frac{\rho*{\sin\left( {180 - \theta} \right)}}{1 - {\rho*{\cos\left( {180 - \theta} \right)}}}}} & (7)\end{matrix}$

Equation (1) can be re-written as Equation (8) as follows:

$\begin{matrix}{{Zo} = {j\frac{1}{2}*{\cot\left\lbrack {B} \right\rbrack}*{\quad\left( {{Zout} - {Zload} + \sqrt{\begin{matrix}\left( {\left( {{Zout} - {Zload}} \right)^{2} -} \right. \\\left. \left( {4*{Zload}*{Zout}*{\tan\left\lbrack {B} \right)}} \right\rbrack^{2} \right)\end{matrix}}} \right)}}} & (8) \\{{where}:} & \; \\{{B} = {\frac{\lambda}{0.25}*\frac{\pi}{2}}} & (9)\end{matrix}$

FIG. 4 is a graphical representation of the trajectories of theimpedance transformations imposed by two example transmission lines136(1) and 136(2) of FIG. 1 having the same VSWR at f0, where the dashedcircle represents all of the locations in the complex impedance planehaving the same VSWR for the impedance ZoComb from the combiner 140. Inthe example of FIG. 4, the two amplifiers 132(1) and 132(2) of FIG. 1have the same output impedance Zout at f0. As such, the two impedancetrajectories start at the same Zout location in the complex impedanceplane. Although the two transmission lines 136(1) and 136(2) havedifferent electrical characteristics at f0 (resulting in two differentimpedance trajectories), those transmission lines are designed such thatthey both end up at two different locations having the same VSWR at f0for the combiner impedance ZoComb, where those two different locationsare both on the real impedance axis (Ω_(Re)). As such, the two amplifiedsignals 138(1) and 138(2) will be combined in phase at the hybrid powercombiner 140 at f0. However, the phase offset at 2f0 will typically notbe appropriate to compensate for the phase-offset deficit at the powercombiner 140.

FIG. 5 is a graphical representation of the trajectories of theimpedance transformations imposed by another two example transmissionlines 136(1) and 136(2) of FIG. 1 having the same VSWR at f0. Here, too,the two amplifiers 132(1) and 132(2) of FIG. 1 have the same outputimpedance Zout at f0, such that the two impedance trajectories againstart at the same Zout location in the complex impedance plane. In thiscase, the two transmission lines 136(1) and 136(2) have differentelectrical characteristics at f0 (resulting in two different impedancetrajectories) that are designed such that they both end up at twodifferent locations having the same VSWR at f0 for the combinerimpedance ZoComb, where neither of those two different locations is onthe real impedance axis (Ω_(Re)). Nevertheless, here, too, the twoamplified signals 138(1) and 138(2) will be combined at the hybrid powercombiner 140 in phase at f0, but typically not 180 degrees out of phaseat 2f0.

FIG. 6 is a graphical representation of the trajectories of theimpedance transformations imposed by two example transmission lines336(1) and 336(2) of FIG. 3. Here, too, the two amplifiers 332(1) and332(2) of FIG. 3 have the same output impedance Zout at f0, such thatthe two impedance trajectories again start at the same Zout location inthe complex impedance plane. In this case, however, the two transmissionlines 336(1) and 336(2) have different electrical characteristics at f0(resulting in two different impedance trajectories) that are designedsuch that they both end up at two different locations having slightlydifferent VSWRs at f0 for the combiner impedance ZoComb, where neitherof those two different locations is on the real impedance axis (Ω_(Re)).

Also shown in FIG. 5 are impedance trajectories for the two transmissionlines 336(1) and 336(2) at the second harmonic frequency 2f0, which endat very different locations on the complex impedance plane. In thisexample, the transmission lines 336(1) and 336(2) have been specificallydesigned such that there will be significant cancellation (i.e.,destructive interference) at the power combiner 340 at 2f0 withoutsignificantly interfering with the phase matching (i.e., constructiveinterference) at f0. In particular, the two transmission lines 336(1)and 336(2) impose a complementary phase offset at 2f0 that issubstantially identical to the 2f0 non-zero phase-offset deficit at thehybrid power combiner 340, while imposing a phase offset at f0 that issubstantially close to zero.

In the example of FIG. 6, the different electrical characteristics ofthe transmission lines 336(1) and 336(2) may result in the twotransmission lines having different power reflection coefficients Γ at2f0. Different power reflection coefficients Γ represent differentequivalent transmission phase angles Ø as given by Equation (7).

Design Example

FIG. 7 is a flow diagram of a technique for implementing thetransmission lines 336(1) and 336(2) of FIG. 3 on a printed circuitboard according to one embodiment of the present disclosure. Thissection describes the flow diagram of FIG. 7 in the context of aparticular design example of the dual-path power amplifier 300 of FIG.3. According to this design example, the impedance Zout from eachamplifier 332 is 27 ohms, and the impedance ZoComb from the powercombiner 340 is 50 ohms. The goal of the technique is to design bothtransmission lines 336(1) and 336(2) to have approximately the samemagnitude ρ of the power reflection coefficient Γ at the fundamentalfrequency f0. The range of values for ρ is from 0 to 1, where ρ=1 is aworst-case scenario where all of the incident signal reaching a load isreflected based to the signal source (i.e., total signal loss). In aperfect system where all of the incident signal goes into the load(i.e., no signal loss), ρ=0. A value of ρ=0.1 corresponds to a practicaland acceptable real-world situation in which about 90% of the incidentsignal goes into the load and only about 10% is reflected back. For thedesign example, it is assumed that ρ=0.1, where the phase offset betweenthe two transmission lines at the second harmonic frequency 2f0 plus thephase offset of the power combiner 340 is substantially close to 180degrees.

In step 702 of FIG. 7, the performance of the hybrid coupler used toimplement the power combiner 340 is characterized at the second harmonicfrequency 2f0. In particular, the phase shift from the power combiner'sfirst input port to the power combiner's output port at 2f0 and thephase shift from the power combiner's second input port to the powercombiner's output port at 2f0 are independently measured. The phaseoffset of the power combiner 340 is the difference between those twomeasured phase shifts. In this design example, the phase offset of thepower combiner 340 is 135 degrees. As such, the goal is to determine theimpedance characteristics of the two transmission paths 336(1) and336(2) that will achieve a complementary phase offset substantiallyequal to 45 degrees at 2f0 such that the total phase offset for the twosignals will be substantially equal to 180 degrees at 2f0 in order forsecond-order harmonic signal energy in those two signals todestructively interfere in the power combiner 340, thereby enabling thegeneration of an amplified output signal 142 having significantlyreduced or negligible harmonic distortion.

In step 704, an amount of phase shift Ø for each transmission line336(1), 336(2) at 2f0 is defined. In this design example, the phaseshift Ø₁ for the first transmission line 336(1) at 2f0 is −39 degrees,and the phase shift Ø₂ for the second transmission line 336(2) at 2f0 is+6 degrees, such that the phase offset between the two transmissionlines at 2f0 will be 45 degrees. Because the phase shift Ø₂ at 2f0 forthe second transmission line 136(2) is only +6 degrees, theimplementation of that second transmission line 136(2) as a PCB metaltrace can be similar to the metal trace 236(2) of the prior-artimplementation 200. As such, only the design of the PCB metal trace forthe implementation of the first transmission line 136(1) having a phaseshift Ø₁ at 2f0 of −39 degrees will need to be optimized to meet thephysical layout constraints on the PCB.

In step 706, the magnitude p of the power reflection coefficient Γ foreach transmission line 336(1), 336(2) at 2f0 is determined. For thisdesign example, the ratio of the power reflection coefficient magnitudeρ₁ for the first transmission line 336(1) to the power reflectioncoefficient magnitude ρ₂ for the second transmission line 336(2) at 2f0is 3.85:1. The ratio 3.85:1 is related to the phase shifts Ø₁ and Ø₂ ofthe first and second transmission lines 136(1) and 136(2) and thedesigned phase offset between them. Furthermore, for this designexample, the power reflection coefficient magnitude ρ₁ for the firsttransmission line 336(1) at 2f0 is 0.64. As understood by those skilledin the art, the value of 0.64 for ρ₁ was selected using a Smith chartplot for microwave amplifier design. As such, given the 3.85:1 ratio,the power reflection coefficient magnitude ρ₂ for the secondtransmission line 336(2) at 2f0 is 0.166.

In step 708, the phase angle θ of the power reflection coefficient Γ foreach transmission line 336(1), 336(2) at 2f0 is determined. UsingEquation (7), given that Ø₁=−39 degrees and that ρ₁=0.64, the phaseangle θ₁ for the power reflection coefficient Γ₁ for the firsttransmission line 336(1) at 2f0 is determined to be −138.3 degrees.Similarly, given that Ø₂=+6 degrees and that ρ₂=0.166, the phase angleθ₂ for the power reflection coefficient Γ₂ for the second transmissionline 336(2) at 2f0 is determined to be +149.2 degrees.

In step 710, an amount of phase shift Ø for each transmission line336(1), 336(2) at the fundamental frequency f0 is defined. In thisdesign example, the phase offset between the two transmission lines atf0 was selected to be 8 degrees, where an 8-degree phase offset at f0corresponds to relatively small impact to the overall power level of theamplified output signal 150 at f0. In particular, in this designexample, the first transmission line 336(1) is designed to have a phaseangle Ø₁ at f0 equal to −3 degrees, and the second transmission line336(2) is designed to have a phase angle θ2 at f0 equal to +5 degrees,such that the phase offset between the two transmission lines will be 8degrees. These phase angles Ø₁ and Ø₂ at f0 are obtained as a directresult of values for the power reflection coefficients Γ₁ and Γ₂ at 2f0.

In step 712, the power reflection coefficients Γ for the first andsecond transmission lines 336(1) and 336(2) at f0 are determined. UsingEquation (7) in an iterative manner, given that the phase angle Ø₁ forthe first transmission line 336(1) at f0 is −3 degrees and with a goalof 0.1 for the power reflection coefficient magnitude ρ₁, the powerreflection coefficient magnitude ρ₁ at f0 is determined to be 0.185, andthe power reflection coefficient phase angle θ₁ at f0 is determined tobe +155.6 degrees. Similarly, given that the phase angle Ø₂ for thesecond transmission line 336(2) at f0 is +5 degrees and with a goal of0.1 for the power reflection coefficient magnitude ρ₂, the powerreflection coefficient magnitude ρ₂ at f0 is determined to be 0.098, andthe power reflection coefficient phase angle θ₂ at f0 is determined tobe −41.01 degrees.

In step 714, the impedances Zload presented to the power combiner 340 byeach transmission line 336(1), 336(2) at f0 and 2f0 are determined.Using Equation (6), given ZoComb=50 ohms and Γ₁(f0)=(ρ₁=0.185, θ₁=+155.6degrees) (in polar coordinates), the impedance Zload for the firsttransmission line 336(1) at f0 (i.e., Zload₁(f0)) is determined to be(35.2+j5.63) ohms. Similarly, given ZoComb=50 ohms and Γ₁(2f0)=(0.641,−138.3 degrees), the impedance Zload₁(2f0) for the first transmissionline 336(1) is determined to be (12.4−j18.0) ohms. Similarly, givenZoComb=50 ohms and Γ₂(f0)=(0.098, −41.01 degrees), the impedanceZload₂(f0) for the second transmission line 336(2) is determined to be(57.5−j7.5) ohms. Similarly, given ZoComb=50 ohms and Γ₂(2f0)=(0.166,+149.2 degrees), the impedance Zload₂(2f0) for the second transmissionline 336(2) is determined to be (37.0+j6.5) ohms.

In step 716, the first and second transmission lines 336(1) and 336(2)are implemented as two metal traces with two lumped capacitors on aprinted circuit board that transform the Zout impedance from theamplifiers 332(1) and 332(2) to the Zload impedances presented to thepower combiner 340 at f0 and 2f0. For this design example, the firsttransmission line 336(1) should be implemented to transform the 27-ohmZout impedance to Zload₁(f0)=(35.2+j5.63) ohms and toZload₁(2f0)=(12.4−j18.0) ohms. Similarly, the second transmission line336(2) should be implemented to transform the 27-ohm Zout impedance toZload₂(f0)=(57.5−j7.5) ohms and to Zload₁ (2f0)=(37.0+j6.5) ohms.

Transmission lines can be implemented in many different ways to achievedesired impedance transformations. For example, a transmission line canbe implemented using a metal trace having one or more serially connectedtrace segments, where each trace segment corresponds to a differentphysical section of the metal trace that performs a different portion ofthe overall impedance transformation of the transmission line. Inaddition, the transmission line can be implemented using one or morelumped elements connected to the metal trace where the one or morelumped elements also perform one or more different portions of theoverall impedance transformation of the transmission line. In thisdesign example, each transmission line 336(1), 336(2) is implemented asa metal trace having three trace segments and one lumped capacitorconnected as a shunt capacitor between the metal trace and a groundterminal.

Using Equations (8) and (9) in an iterative manner, given Zout=27 ohmsand the two different Zload₁ values at f0 and 2f0, the differentportions of the overall impedance transformations corresponding to thethree trace segments and the lumped capacitor for the first transmissionline 336(1) can be determined. Similarly, given Zout=27 ohms and the twodifferent Zload₂ values at f0 and 2f0, the different portions of theoverall impedance transformations corresponding to the three tracesegments and the lumped capacitor for the second transmission line336(2) can also be determined.

Table I presents the results of this iterative processing. In Table I,Zbegin refers to the impedance at the beginning of a portion of theoverall impedance transformation for a transmission line at either f0 or2f0, and Zend refers to the impedance at the end of that portion. Thelength λ is the number of wavelengths corresponding to that portion ateither f0 or 2f0, and Zo is the impedance of that portion. For example,the first entry in Table I corresponds to the first trace segment of thefirst transmission line 336(1) at f19, where that first trace segmenthas a beginning impedance Zbegin of (35.2+j5.63) ohms at f0 and anending impedance Zend of (31.2+j12.1) at f0. To achieve that portion ofthe overall impedance transformation of the first transmission line336(1) at f0, the first trace segment has a length λ of 0.013wavelengths and an impedance Zo of 11.8 ohms.

TABLE I Tx Line Freq Portion Zbegin (ohms) Zend (ohms) Length (λ) ZoFirst f0 1^(st) Segmt  35.2 + j5.63  31.2 + j12.1 0.013 11.8 ohms Firstf0 2^(nd) Segmt  31.2 + j12.1 20.0 − j0.0 0.152 29.9 ohms First f03^(rd) Segmt 20.0 − j0.0 22.3 − j9.7 0.055 39.5 ohms First f0 Capacitor22.3 − j9.7 27 — 2.7 pF First 2f0 1^(st) Segmt  12.4 − j18.0  21.2 −j21.5 0.026 11.8 ohms First 2f0 2^(nd) Segmt  21.2 − j21.5  14.0 + j11.60.304 29.9 ohms First 2f0 3^(rd) Segmt  14.0 + j11.6  14.5 − j13.3 0.11039.5 ohms First 2f0 Capacitor  14.5 − j13.3 27 — 2.7 pF Second f0 1^(st)Segmt 57.5 − j7.5 57.2 + j8.8 0.030 29.2 ohms Second f0 2^(nd) Segmt57.2 + j8.8 53.4 + j8.6 0.035 55.0 ohms Second f0 3^(rd) Segmt 53.4 +j8.6   27 − j0.8  0.2225 38.9 ohms Second f0 Capacitor   27 − j0.8 27 —0.2 pF Second 2f0 1^(st) Segmt   37 + j6.5 29.6 + j11  0.060 29.2 ohmsSecond 2f0 2^(nd) Segmt 29.6 + j11  29.5 − j8.6 0.070 55.0 ohms Second2f0 3^(rd) Segmt 29.5 − j8.6 26.8 − j1.9 0.445 38.9 ohms Second 2f0Capacitor 26.8 − j1.9 27 — 0.2 pF

Note that, in Table I, the different portions of each overall impedancetransformation are described starting from the corresponding input ofthe power combiner 340 and progressing back to the output of thecorresponding amplifier 332(1), 332(2). Thus, the impedance at thebeginning of the first trace segment of the first transmission line336(1) at f0 is the impedance at the first input of the power combiner140 at f0, and the impedance at the “end” of the lumped capacitor is theimpedance at the output of the first amplifier 332(1) (e.g., 27 ohms).As such, as reflected in Table I, the impedance at the end of the firsttrace segment of the first transmission line 336(1) at f0 is the same asthe impedance at the beginning of the second trace segment of the firsttransmission line 336(1) at f0. Similarly, the impedance at the end ofthe second trace segment of the first transmission line 336(1) at f0 isthe same as the impedance at the beginning of the third trace segment ofthe first transmission line 336(1) at f0, and the impedance at the endof the third trace segment of the first transmission line 336(1) at f0is the same as the impedance at the “beginning” of the lumped capacitorat f0. These same characteristics apply to the first transmission line336(1) at 2f0 and to the second transmission line 336(2) at both f0 and2f0.

Since impedances of series-connected elements are sequentially additive,the three different trace segments and the lumped capacitor of Table Iare implemented sequentially as described to achieve the desiredimpedance transformation. Other implementations may involve differentsequences of trace segments and/or lumped elements.

For PCB-based implementations in which the transmission lines 336(1) and336(2) are implemented using two metal traces and lumped elements on aPCB, the design strategy of FIG. 7 involves purposely designing the twotransmission lines to provide different impedance transformations at thesecond harmonic frequency 2f0 in order to achieve different amounts ofphase shift at 2f0 in the two different amplified signals that arrive atthe power combiner 340, such that the resulting output signal 342 willhave reduced distortion at the second harmonic frequency withoutsignificantly impacting the power level of the output signal at thefundamental frequency. In typical implementations of this designstrategy, the two different metal traces will have different physicalconfigurations (e.g., lengths, widths, shapes) that are not identical(e.g., not mirror images of each other).

Those skilled in the art understand that a trace segment can be designedin different ways to achieve a given impedance, where, in general,impedance is a function of length, thickness, width, and materialcomposition of the trace segment. The design of a metal trace may beimpacted by external constraints such as the available footprint on thePCB and the particular metal to be used.

FIG. 8 is a simplified, top view of a PCB-based implementation 800 ofthe dual-path amplifier 300 of FIG. 3, according to certain embodimentsof the present disclosure. As shown in FIG. 8 and similar to theconventional PCB-based implementation 200 of FIG. 2, the power splitter320 and the power combiner 340 of FIG. 3 may be respectively implementedusing two symmetric instances 820 and 840 of the same discrete, hybridcoupler that are mounted onto the PCB 805. Also mounted onto the PCB 805is a packaged IC device 832 that contains the two amplifiers 332(1) and332(2) of FIG. 3. In an alternative embodiment, each amplifier 332(1),332(2) may be implemented in a separate packaged IC device. In onepossible implementation, the hybrid power splitter 820 and the hybridpower combiner 840 are two instances of an X3C09P1-03S 3 dB 90-degXinger Hybrid Coupler 800-1000 MHz from Anaren of East Syracuse, N.Y.,and the packaged IC device 832 is an A2I09VD050 RF Amplifier from NXPSemiconductors of Eindhoven, Netherlands.

As shown in FIG. 8, the two transmission lines 336(1) and 336(2) of FIG.3 are implemented using two respective metal traces 836(1) and 836(2) onthe PCB 805. Note that, in this case and unlike the situation in theconventional implementation 200 of FIG. 2, the physical layouts of thetwo metal traces 836(1) and 836(2) are intentionally designed not to beidentical (e.g., not to be mirror images of each other), such that thetwo transmission lines 336(1) and 336(2) will have different electricalcharacteristics at the second harmonic frequency 2f0 such that the two,second harmonic signal components 338(1) and 338(2) of FIG. 3 willarrive at the hybrid power combiner 840 with a complementary phaseoffset at 2f0 that substantially compensates for the non-zerophase-offset deficit of the hybrid power combiner 840. In particular, inthis example, the metal trace 836(2) has certain trace segments that arewider than the corresponding trace segments of the metal trace 836(1) Inaddition, the two metal traces 836(1) and 836(2) may be designed suchthat the transmission lines 336(1) and 336(2) will have sufficientlysimilar electrical characteristics at the fundamental frequency f0 suchthat there will be minimal power loss and distortion in the outputsignal 342 at the output node 350 of the power amplifier 300 at f0. Inparticular, the goal is to design the two metal traces 836(1) and 836(2)to have different and independent electrical characteristics that do notintroduce significant losses at the fundamental frequency f0 but dointroduce a desired phase offset between the two signal components338(1) and 338(2) at the second harmonic frequency 2f0.

The total RF source signal phase offset at 2f0 between the twotransmission lines 336(1) and 336(2) at the combining port referenceplane (i.e., inside the hybrid power combiner 840) should be close to180 degrees out of phase to achieve significant secondary harmoniccancellation. Due to physical dimensions limitations, the power combiner840 fails to provide enough phase offset to achieve the full required180 degrees. The proposed design takes advantage of the phase offsetthat is available and inherent in the power combiner 840 at 2f0 andcomplements that phase offset with additional phase offset coming fromthe designed transmission lines 336(1) and 336(2). The resultantaccumulated total phase offset at 2f0 will be sufficient to inducesignificant secondary harmonic cancellation without significantlyimpacting the power of the amplified output signal at f0.

According to an embodiment, the different electrical characteristics ofthe two transmission lines 336(1) and 336(2) are implemented bydesigning the metal traces 836(1) and 836(2) with different physical andelectrical characteristics. For example, each of the metal traces 836(1)and 836(2) has a first end (input end) coupled to an amplifier output,and a second end (output end) coupled to an input port of the powercombiner 840, where each metal trace is made up of metal trace segmentswhose physical characteristics (e.g., length, width, shape, material)are selected to achieve desired electrical characteristics. As anexample, all other things being equal, doubling the width of one metaltrace relative to the width of the other metal trace will result in thewider trace having an impedance that is half the impedance of the othertrace, which will in turn increase the phase offset between the twotraces at 2f0.

Although embodiments have been described in the context of the secondharmonic frequency, in general, embodiments of the present disclosurecan be designed for other harmonic frequencies in addition to or insteadof the second harmonic frequency.

According to certain embodiments, disclosed is a dual-path amplifierconfigured to operate at a fundamental frequency of operation. Thedual-path amplifier comprises a power splitter, first and second poweramplifiers, first and second transmission lines, and a power combiner.The power splitter is configured to split an input signal into first andsecond signal components. The first power amplifier is configured toamplify the first signal component to generate a first amplified signalcomponent, and the second power amplifier is configured to amplify thesecond signal component to generate a second amplified signal component.The first transmission line is configured to receive the first amplifiedsignal component and output a first transformed amplified signalcomponent, and the second transmission line is configured to receive thesecond amplified signal component and output a second transformedamplified signal component. The power combiner is configured to combinethe first and second transformed amplified signal components to generatean amplified output signal. The first and second transmission lines areconfigured to compensate for a non-zero phase-offset deficit in thepower combiner at a harmonic of the fundamental frequency such that thefirst and second transmission lines and the power combiner reducedistortion in the amplified output signal at the harmonic frequency.

According to certain embodiments, disclosed is a method of amplifying aninput signal to generate an amplified output signal. The methodcomprises splitting the input signal into first and second signalcomponents; amplifying the first signal component to generate a firstamplified signal component; amplifying the second signal component togenerate a second amplified signal component; transmitting the firstamplified signal component along a first transmission line to output afirst transformed amplified signal component; transmitting the secondamplified signal component along a second transmission line to output asecond transformed amplified signal component; and combining the firstand second transformed amplified signal components at a power combinerto generate an amplified output signal. The first and secondtransmission lines compensate for a non-zero phase-offset deficit in thepower combiner at a harmonic frequency of a fundamental frequency suchthat the first and second transmission lines and the power combinerreduce distortion in the amplified output signal at the harmonicfrequency.

The preceding detailed description is merely illustrative in nature andis not intended to limit the embodiments of the subject matter or theapplication and uses of such embodiments. As used herein, the word“exemplary” means “serving as an example, instance, or illustration.”Any implementation described herein as exemplary is not necessarily tobe construed as preferred or advantageous over other implementations.Furthermore, there is no intention to be bound by any expressed orimplied theory presented in the preceding technical field, background,or detailed description.

The connecting lines shown in the various figures contained herein areintended to represent exemplary functional relationships and/or physicalcouplings between the various elements. It should be noted that manyalternative or additional functional relationships or physicalconnections may be present in an embodiment of the subject matter. Inaddition, certain terminology may also be used herein for the purpose ofreference only, and thus are not intended to be limiting, and the terms“first”, “second” and other such numerical terms referring to structuresdo not imply a sequence or order unless clearly indicated by thecontext.

As used herein, a “node” means any internal or external reference point,connection point, junction, signal line, conductive element, or thelike, at which a given signal, logic level, voltage, data pattern,current, or quantity is present. Furthermore, two or more nodes may berealized by one physical element (and two or more signals can bemultiplexed, modulated, or otherwise distinguished even though receivedor output at a common node).

The foregoing description refers to elements or nodes or features being“connected” or “coupled” together. As used herein, unless expresslystated otherwise, “connected” means that one element is directly joinedto (or directly communicates with) another element, and not necessarilymechanically. Likewise, unless expressly stated otherwise, “coupled”means that one element is directly or indirectly joined to (or directlyor indirectly communicates with, electrically or otherwise) anotherelement, and not necessarily mechanically. Thus, although the schematicshown in the figures depict one exemplary arrangement of elements,additional intervening elements, devices, features, or components may bepresent in an embodiment of the depicted subject matter.

While at least one exemplary embodiment has been presented in theforegoing detailed description, it should be appreciated that a vastnumber of variations exist. It should also be appreciated that theexemplary embodiment or embodiments described herein are not intended tolimit the scope, applicability, or configuration of the claimed subjectmatter in any way. Rather, the foregoing detailed description willprovide those skilled in the art with a convenient road map forimplementing the described embodiment or embodiments. It should beunderstood that various changes can be made in the function andarrangement of elements without departing from the scope defined by theclaims, which includes known equivalents and foreseeable equivalents atthe time of filing this patent application.

What is claimed is:
 1. A dual-path amplifier configured to operate at afundamental frequency of operation, the dual-path amplifier comprising:a power splitter configured to split an input signal into first andsecond signal components; a first power amplifier configured to amplifythe first signal component to generate a first amplified signalcomponent; a second power amplifier configured to amplify the secondsignal component to generate a second amplified signal component; afirst transmission line configured to receive the first amplified signalcomponent and output a first transformed amplified signal component; asecond transmission line configured to receive the second amplifiedsignal component and output a second transformed amplified signalcomponent; and a power combiner configured to combine the first andsecond transformed amplified signal components to generate an amplifiedoutput signal, wherein: the first and second transmission lines areconfigured to compensate for a non-zero phase-offset deficit in thepower combiner at a harmonic of the fundamental frequency such that thefirst and second transmission lines and the power combiner reducedistortion in the amplified output signal at the harmonic frequency. 2.The amplifier of claim 1, wherein: the harmonic of the fundamentalfrequency is a second harmonic frequency; the power combiner is a hybridpower combiner that applies a first phase offset substantially equal to90 degrees to signal components at the fundamental frequency and asecond phase offset that is substantially less than 180 degrees tosignal components at the second harmonic frequency, wherein thephase-offset deficit of the power combiner is equal to 180 degrees minusthe second phase offset at the second harmonic frequency; and the firstand second transmission lines are configured to provide a complementaryphase offset substantially equal to the phase-offset deficit of thepower combiner.
 3. The amplifier of claim 2, wherein the second phaseoffset at the second harmonic frequency is no more than about 80degrees.
 4. The amplifier of claim 1, wherein the first and secondtransmission lines comprise respective first and second metal traces ona printed circuit board (PCB).
 5. The amplifier of claim 4, wherein: thefirst transmission line imposes a first impedance transformation at theharmonic frequency that results in a first phase shift at the harmonicfrequency; and the second transmission line imposes a second impedancetransformation at the harmonic frequency that results in a second phaseshift at the harmonic frequency, wherein a non-zero difference betweenthe first and second phase shifts at the harmonic frequencysubstantially compensates for the non-zero phase-offset deficit of thepower combiner.
 6. The amplifier of claim 4, wherein that first andsecond metal traces have different physical characteristics that resultin different electrical characteristics
 7. The amplifier of claim 6,wherein the first and second metal traces are not mirror images of eachother.
 8. The amplifier of claim 4, wherein the first and secondtransmission lines each further comprise one or more lumped elementsmounted on the PCB and connected to the corresponding metal trace.
 9. Amethod of amplifying an input signal to generate an amplified outputsignal, the method comprising: splitting the input signal into first andsecond signal components; amplifying the first signal component togenerate a first amplified signal component; amplifying the secondsignal component to generate a second amplified signal component;transmitting the first amplified signal component along a firsttransmission line to output a first transformed amplified signalcomponent; transmitting the second amplified signal component along asecond transmission line to output a second transformed amplified signalcomponent; and combining the first and second transformed amplifiedsignal components at a power combiner to generate an amplified outputsignal, wherein: the first and second transmission lines compensate fora non-zero phase-offset deficit in the power combiner at a harmonicfrequency of a fundamental frequency such that the first and secondtransmission lines and the power combiner reduce distortion in theamplified output signal at the harmonic frequency.
 10. The method ofclaim 9, wherein: the harmonic of the fundamental frequency is a secondharmonic frequency; the power combiner is a hybrid power combiner thatapplies a first phase offset substantially equal to 90 degrees to signalcomponents at the fundamental frequency and a second phase offset thatis substantially less than 180 degrees to signal components at thesecond harmonic frequency, wherein the phase-offset deficit of the powercombiner is equal to 180 degrees minus the second phase offset at thesecond harmonic frequency; and the first and second transmission linesprovide a complementary phase offset substantially equal to thephase-offset deficit of the power combiner.
 11. The method of claim 10,wherein the second phase offset at the second harmonic frequency is lessthan about 80 degrees.
 12. The method of claim 9, wherein the first andsecond transmission lines comprise respective first and second metaltraces on a PCB.
 13. The method of claim 12, wherein: the firsttransmission line imposes a first impedance transformation at theharmonic frequency that results in a first phase shift at the harmonicfrequency; and the second transmission line imposes a second impedancetransformation at the harmonic frequency that results in a second phaseshift at the harmonic frequency, wherein a non-zero difference betweenthe first and second phase shifts at the harmonic frequencysubstantially compensates for the non-zero phase-offset deficit of thepower combiner.
 14. The method of claim 12, wherein that first andsecond metal traces have different physical characteristics resulting indifferent electrical characteristics.
 15. The method of claim 14,wherein the first and second metal traces are not mirror images of eachother.
 16. The amplifier of claim 12, wherein the first and secondtransmission lines each further comprise one or more lumped elementsmounted on the PCB and connected to the corresponding metal trace.